Theory of Operation—492/492P Service Vol. 1 (SN B030000 & up)
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C2077. Transformer T1077 serves as the amplifier feedback
path and also as part of the oscillator resonator. Tuning
varactor CR1075 provides virtually all of the resonator ca
pacitance to allow a wide tuning range. Transformers T1077
and T1075 constitute the resonator inductance that may be
selected by using different tap combinations to interconnect
the two coils. Resonator inductance is adjusted to center
the oscillator's tune range near 18 MHz. R2076 and R2079
enhance the amplifier high frequency stability.
A discrete two-stage amplifier provides an unsaturated
voltage gain of approximately 43 dB for the difference fre
quency signal from the 2200 MHz Reference Mixer. The
output of Q1036 drives a differential amplifier consisting of
Q1037 and Q1038. The differential stage limits the output
swing to ECL compatible levels. Dc bias for the amplifier is
provided by Q1036, which has dc collector-base feedback
via voltage divider R1039 and R1041. ECL line receivers
U2036D and U2036B buffer signals from the discrete ampli
fier and the 14-22 MHz oscillator, respectively. Output sig
nals from
these
amplifiers
are
applied
to
the
phase/frequency detector for comparison.
A pair of ECL D-type flip-flops (U2027A, U2027B) com
prise the phase/frequency detector. The flip-flop outputs are
wired and connected to the input of U2036C, which serves
as a common reset. The clock input to U2027B is the 14-
22 MHz VCO signal, and the clock input to U2027A is the
amplified difference signal from the Reference Mixer. If the
two clock signals are of identical phase and frequency, both
flip-flop sections set then reset together. If the phase of the
Reference Mixer signal leads the 14-22 MHz signal,
U2027A will remain set longer than U2027B. If the signal
lags, U2027B will set first and remain set longer. The signal
that leads in phase or has the higher frequency will cause
the associated flip-flop to remain set a higher percentage of
the time. The average differential output voltage of the two
flip-flops therefore indicates whether the Reference Mixer
signal leads, lags, or differs in frequency from the 14-
22 MHz VCO reference. Output of the detector is filtered by
an RC lowpass filter, then applied to differential amplifier
U1028, which tunes the 2182 MHz oscillator.
The phaselock circuit adjusts the Microstrip Oscillator
frequency such that the Reference Mixer output always
matches the frequency of the 14-22 MHz VCO. The
Microstrip Oscillator is therefore locked to a frequency equal
to that of the 2200 MHz reference minus that of the 14-
22 MHz VCO. If the 14-22 MHz Oscillator is swept or tuned,
the Microstrip Oscillator sweeps and tunes an equal
amount. Within the control bandwidth of the lock loop, the
Microstrip Oscillator FM noise is reduced to that of the ref
erence circuitry. The phaselock loop bandwidth is controlled
by R1024, C1026, and R1025, C1023. Unity gain for the
phaselock loop occurs near 200 kHz with a gain slope of
—6 dB/octave. The gain slope breaks to —12 dB/octave for
frequencies below 16 kHz. Resistors R1030 and R1031 di
vide and offset the output of U1028 so the Microstrip oscil
lator tune voltage ranges between 0 and —12.5 V.
CAVITY 2ND LOCAL OSCILLATOR <38
Refer to the block diagram adjacent to Diagram 38. The
Cavity 2nd Local Oscillator generates the 2182 MHz signal
that is:
1) mixed with the 2072 MHz signal from the 1st Converter
to produce the 110 MHz intermediate frequency in the
2072 MHz 2nd Converter; and
2) used as a reference in the harmonic mixer in the phase
lock circuit of the 829 MHz 2nd Converter.
The oscillator is a low noise cavity oscillator that free-
runs at a nominal frequency of 2182 MHz, but is tunable
over a range of 8 MHz. A relatively large resonant cavity
with very high Q allows the oscillator to operate at low noise
levels and with a power output of +10 dBm. The cavity
itself operates in the TEM mode and utilizes a foreshortened
vertical post to form a coaxial structure.
Two equivalent schematic diagrams are shown in Fig.
5-3, a direct connection representation, and the RF
equivalent.
As shown in the RF equivalent diagram, transistor Q1
operates as a common emitter oscillator with positive
feedback in the collector circuit. It is biased to operate with
an emitter current of approximately 30 mA. The collector is
coupled to the tunable resonant cavity by a coupling screw.
Line length between the transistor collector and the coupling
screw is set by an adjustable wire strap.
Energy distribution inside the cavity is such that E fields
are at the top of the cavity and magnetic (H) fields circulate
at the bottom. Energy is extracted from the tank circuit (cav
ity) by inductive coupling near the bottom of the cavity. The
output connectors, with attached coupling loops, are
rotated to adjust the power output level from the oscillator.
One connector is adjusted to provide +10 dBm of output,
and the other is set to provide 0 dBm of output power.
Tuning of the oscillator frequency is by means of a
varactor diode that is controlled by a 15 to 40 V bias signal
from the Shaper and Bias circuit. This signal varies the oscil
lator frequency over an 8 MHz range. The diode is located
near the top of the cavity and is coupled to the cavity post
by a capacitive coupling hat (E-field coupling). RF energy in
the coupling hat is decoupled from the varactor bias
feedthrough by an inductor. The spacing between the hat
and the post determines the sensitivity for the diode tuning.
5-12
R EV A U G 1981
